THAT2252 RMS Detector Replacement Using A THAT300 Array

[mediatechnology]

Not to quibble but the 2252 is a RMS (cough) convertor so there is a 2xVbe term to perform the X^2 operation, minus 1Vbe for square root. (there is also an integration operation not shown. )

The TS-1 is just a simple LOG conversion to extract relative dB levels. I've done plenty RMS conversions with that array but it would have more going on (integration and square root involving all the transistors). In those the full wave rectifier is done more conventionally before the RMS operation.

Of course.
I'm talking about the absolute value portion only.
(Which is the toughest part to do well.)

I grafted your TS-1 absolute value (similar to OA1 and associated Qs) onto the 2252's logX2 current input at OA2's inverting input.
You may have missed that statement.
The TS-1's absolute value is what I found to be useful.
If you compare the absolute value portion of the circuits they are nearly identical.

Since the detector will be stereo and true power summed I can use 2X THAT300 transistors for the current mirror and re-use the other two in the opposite channel.
One THAT300 will be Q4-Q6 plus Q7 (for V3). (4 Q total.)
Half a THAT300 for Q1 and Q2.
Someone not wanting to use half a THAT300 for the current mirror in SMT could use a NST45011 dual.

Total 3X THAT300 plus 3 dual op amps for a stereo True Power Summed detector.
Not as efficient as a pair of 2252s but using available parts.

[JR.]

Not to quibble but the 2252 is a RMS (cough) convertor so there is a 2xVbe term to perform the X^2 operation, minus 1Vbe for square root. (there is also an integration operation not shown. )

The TS-1 is just a simple LOG conversion to extract relative dB levels. I've done plenty RMS conversions with that array but it would have more going on (integration and square root involving all the transistors). In those the full wave rectifier is done more conventionally before the RMS operation.

Of course.
I'm talking about the absolute value portion only.
(Which is the toughest part to do well.)

I grafted your TS-1 absolute value (similar to OA1 and associated Qs) onto the 2252's logX2 current input at OA2's inverting input.
You may have missed that statement.
The TS-1's absolute value is what I found to be useful.
If you compare the absolute value portion of the circuits they are nearly identical.

Great minds think alike? Something similar (but not exact) was done inside ne570-572 rectifiers. In fact I did a compressor for Loft (Phoenix Audio Lab) where I used ne572s just for the two rectifiers inside each chip. Back then PCB real estate could be an issue for a complex designs and I needed multiple rectifiers for comp limiter and de-essing. From memory the ne570 type rectifiers were reasonable over a roughly 70 dB range (maybe?).

Since the detector will be stereo and true power summed I can use 2X THAT300 transistors for the current mirror and re-use the other two in the opposite channel.
One THAT300 will be Q4-Q6 plus Q7 (for V3). (4 Q total.)
Half a THAT300 for Q1 and Q2.
Someone not wanting to use half a THAT300 for the current mirror in SMT could use a NST45011 dual.

Total 3X THAT300 plus 3 dual op amps for a stereo True Power Summed detector.
Not as efficient as a pair of 2252s but using available parts.

For chuckles you might compare with added emitter degeneration resistors, but I'm not even sure how to measure that error term? If you try to measure using DC the input offset could become significant.

JR

[mediatechnology]

Great minds think alike?

I think so.
BTW John we had a link to the 570-based Compander Noise Reduction system you (Phoenix) did that was hosted on the Circular Science site.
I noticed that link was broken.
If you want to send it to me I'll host it.

I'll draw up the current-rectified version of the dbx2252.

This is the NE570 rectifier:


NE570 Rectifier Simplified Diagram

Q6, a PNP is inside the feedback loop of the op amp formed by Q1-Q4.
Q5 and Q7 are the current mirror.
Q7 is outside the feedback loop.

Related reading:
Signetics NE570 NE571/572 Compander Product Guide https://www.proaudiodesignforum.com/for ... 407&p=4571
Thanks for sending this one to me John.

[JR.]

Great minds think alike?

I think so.
BTW John we had a link to the 570-based Compander Noise Reduction system you (Phoenix) did that was hosted on the Circular Science site.
I noticed that link was broken.
If you want to send it to me I'll host it.

Yup my vanity website is in bad shape,,, I guess I'm not that vain?

I did a 570 based NR in 1977 that was stock as a stone single NE570

Later I did a 572 based NR (P-522) that almost had more tricks in it than the TS-1

If I was ever going to teach an engineering design class, I'd show them the TS-1 schematic, and this one and ask them to explain what every part is doing.

p-522.jpg (188.66 KiB) Viewed 12653 times

I'll draw up the current-rectified version of the dbx2252.

This is the NE570 rectifier:


NE570 Rectifier Simplified Diagram

Q6, a PNP is inside the feedback loop of the op amp formed by Q1-Q4.
Q5 and Q7 are the current mirror.
Q7 is outside the feedback loop.

I guess one of Q7's two collectors is kind of inside the loop. My first exposure to compound devices was in the early '70s working with Interdesign a technology to make semi-custom ICs... they used a standard IC with everything but the top metallization layer, and by adding a custom metallization layer connection pattern you could reconfigure the many devices inside to do different tricks... pretty clever for '70s. While working at that pitch shift company we made a semi custom chip set to do tricks using that technology.

Related reading:
Signetics NE570 NE571/572 Compander Product Guide viewtopic.php?f=12&t=407&p=4571
Thanks for sending this one to me John.

Looking at figure 6 I'm not sure Cr was inside the chip, the effective C was a few uF so external. Perhaps not intended, but I connected that node (in a 572) to the virtual earth - input of an op amp to convert back to a voltage. You could do a log conversion just by hanging a diode junction to ground there.

JR

[mediatechnology]

Q7 is drawn weird even for a simplified drawing.
I don't see a second collector but do see two bases which is also weird.

Looking at Q3 which is drawn the same way do you suppose that was the draftsman's shorthand for a two transistor current mirror?

Cr is off-chip but is shown without a terminal.

What's also not shown correctly but is stated in the text is that Q1's base is actually at the 1.8V reference which places the summing node at 1.8V.

[JR.]

Q7 is drawn weird even for a simplified drawing.
I don't see a second collector but do see two bases which is also weird.

q7 has one base coming out at right angle. Second collector is angled down at similar angle to first collector. The second collector is connected to the one base... to form the current mirror. Imagine if Q7 was two identical PNPs with emitters and basses connected together. One collector ties to the bases, the other collector is mirror output

Looking at Q3 which is drawn the same way do you suppose that was the draftsman's shorthand for a two transistor current mirror?

yes Q3 and Q7 are both dual collector devices configured as current mirrors.

Cr is off-chip but is shown without a terminal.

What's also not shown correctly but is stated in the text is that Q1's base is actually at the 1.8V reference which places the summing node at 1.8V.

Yes...

JR

[mediatechnology]

Thanks John for uploading the compander.
I'll go find that broken link in the OP and fix it.

I got out my crayons and sketched the schematics of the various test circuits.

RMS Detector

The first schematic is the 2252-style clone sent to me by Gary Hebert.
The 2252, unlike the original 303, performs absolute value ahead of the RMS stage.
The following schematic is the RMS portion after the absolute value rectifier.


A THAT2252-style RMS detector made from op amps and a THAT300 transistor array.

The input is a current.
Q1 and Q2 provide a 2Xlog output for negative input currents.
D1 is a reverse polarity clamp.
Q3 Rt and Ct are the log-domain filter.
Q4, op amp B and the 1MΩ current source provide level shift.
All transistors should be matched and in thermal equilibrium so a THAT300 array is used.

Rt is not true current source but for practical purposes can be thought as one since the delta-V at the emitter of Q3 is quite small.
Both resistors should be equal value and set the timing current which is √(It1*It2).

The Absolute Value Circuit

The absolute value circuit ahead of the RMS detector can be done a number of ways and can be either voltage or current output.
Since the 2X logging stage uses a current input, a current output is preferred.

The first absolute value stage is a conventional fullwave rectifier:


A Diode-based fullwave rectifier with current output.

Halfwave rectification for positive inputs is performed by D2.
D1 clamps the op amp for negative inputs.
The sum of the currents at the output represent the absolute value of the input voltage.
The negative-going current output of the absolute value can connect directly to the current input of the RMS stage.

The second example shows current rectification as it was originally done in the THAT2252 and Loftec TS-1.

The actual circuit is the TS-1's rectifier. The 2252 is almost exactly the same.


A current rectifier based on the Loftec TS-1 and THAT2252.

I've redrawn the TS-1's rectifier in a "reinterpretation" that I can more easily follow.
I shifted the focus of the current mirror stage, Q3 and Q4 so that Q3 looks more like a rectifier.

Q1, a common base stage provides additional non-inverting voltage gain as a "helper" to improve bandwidth of op amp A.
Q1's collector load is 15K+470R.
The base is biased to -7.5V.
The emitter resistor is 470R making the added inside the loop gain about 30X.

The op amp inputs are held at -3.75V by virtue of the voltage divider formed by the 30K and 10K resistors.
This permits Q2 and Q4 to sink current for loads held at or near ground.

Negative inputs are rectified by the base-emitter junction of Q2.
This sets up a nearly-identical collector current in Q2 which provides the negative polarity rectified output.
The collector current for Q2 (when loaded by the following stage) flows from the output to the inverting input held at -3.75V.
The 470Ω base resistor pre-biases Q2 to reduce rectification deadband.

For positive inputs, the output of op amp A and the collector of Q1 swing more negative.
The emitter base junction of Q3, a diode-connected transistor, provides rectification for positive inputs.
The Q3/Q4 stage is where I took "artistic liberty" in the schematic layout to emphasize Q3's base-emitter junction's roll as rectifier.
Q3 and Q4 are a current mirror with Q3 also a rectifier diode.
As the current in Q3's base emitter junction increases a nearly-identical mirrored current is setup in the collector of Q4.
The action of the current mirror serves to invert polarity for positive inputs.

When the collector currents of Q2 and Q4 are summed, the final result provides the absolute value.
The output of the current rectifier is a current that can be directly connected to the RMS stage.

Q3 and Q4 require matched devices in thermal equilibrium.
For a stereo detector, a single THAT300 could be used and share between channels.
A low-cost dual NPN transistor pair such as the NST45011 could also be used.

Advantages and Disadvantages.

The diode-base absolute value circuit has about 10 dB or so less dynamic range but does not require matched transistors in the rectifier.
The current rectifier has more dynamic range but requires matched devices.

I may have missed a few subtleties but really wanted to get this posted before this session times out. LOL.

Related reading:
A Discussion About True Power Summing for Stereo Compressors https://www.proaudiodesignforum.com/for ... p?f=6&t=61
Compressor Attack Release Signature Comparisons https://www.proaudiodesignforum.com/for ... ?f=6&t=281

[JR.]

Thanks John for uploading the compander.
I'll go find that broken link in the OP and fix it.

I got out my crayons and sketched the schematics of the various test circuits.

RMS Detector

The first schematic is the 2252-style clone sent to me by Gary Hebert.
The 2252, unlike the original 303, performs absolute value ahead of the RMS stage.
The following schematic is the RMS portion after the absolute value rectifier.


A THAT2252-style RMS detector made from op amps and a THAT300 transistor array.

The input is a current.
Q1 and Q2 provide a 2Xlog output for negative input currents.
D1 is a reverse polarity clamp.
Q3 Rt and Ct are the log-domain filter.
Q4, op amp B and the 1MΩ current source provide level shift.
All transistors should be matched and in thermal equilibrium so a THAT300 array is used.

While that looks like the 2252 it lacks some typical RMS aspects. The classic RMS math is square root of the integral of signal squared.

The two Vbe drops in series performs 2xLOG which is equivalent to signal squared when compared to a single Vbe at the decode side. That squared current in Q3 shows up in the faster charging of integration cap Ct, but to finish the RMS equation a square root needs to be performed. This is generally accomplished by biasing the second Vbe (or the first) with the integrated squared current. Both diode drops on the decode side appear to be fixed currents, no doubt representing 0dB at equilibrium.

I won't be a purist or pedant about true or real RMS vs ave because my development work while developing my last peak/VU console meters revealed very little difference (I couldn't see any on my bench) between RMS (actually computed by my microprocessor) and simple average for complex audio signals. (I ended up removing the RMS code because it did not appear to add any value other than perhaps for marketing)

Rt is not true current source but for practical purposes can be thought as one since the delta-V at the emitter of Q3 is quite small.
Both resistors should be equal value and set the timing current which is √(It1*It2).

RT sets the 0dB current, for 0V output.

The Absolute Value Circuit

The absolute value circuit ahead of the RMS detector can be done a number of ways and can be either voltage or current output.
Since the 2X logging stage uses a current input, a current output is preferred.

The first absolute value stage is a conventional fullwave rectifier:


A Diode-based fullwave rectifier with current output.

Halfwave rectification for positive inputs is performed by D2.
D1 clamps the op amp for negative inputs.
The sum of the currents at the output represent the absolute value of the input voltage.
The negative-going current output of the absolute value can connect directly to the current input of the RMS stage.

A subtle issue with the simple diode FW rectifier, is that the op amp DC offset between that op amp and logging op amps can cause a low level error. If the DC errors are in the same direction they partially cancel, but opposite direction Dc offsets add. Back in the day using TL07x op amps with poor DC specs could be a factor.

The second example shows current rectification as it was originally done in the THAT2252 and Loftec TS-1.

The actual circuit is the TS-1's rectifier. The 2252 is almost exactly the same.


A current rectifier based on the Loftec TS-1 and THAT2252.

I've redrawn the TS-1's rectifier in a "reinterpretation" that I can more easily follow.
I shifted the focus of the current mirror stage so that half of it looks more like a rectifier.

Q1, a common base stage provides additional non-inverting voltage gain for op amp A.
It's collector load is 15K+470R.
The base is biased to -7.5V.
The emitter resistor is 470R making the added Inside the loop gain about 30X.

The op amp inputs are held at -3.75V by virtue of the voltage divider formed by the 30K and 10K resistors.
This permits Q2 and Q4 to sink current for loads held at or near ground.

Negative inputs are rectified by the base-emitter junction of Q2.
This sets up a nearly-identical collector current in Q2 which provides the negative polarity rectified output.
The collector current for Q2 (when loaded by the following stage) flows from the output to the inverting input held at -3.75V.
The 470Ω base resistor pre-biases Q2 to reduce rectification deadband.

For positive inputs the output of op amp A and the collector of Q1 swing more negative.
The emitter base junction of Q3 provides rectification for positive inputs.
It is in this stage where I took "artistic liberty" in the schematic layout to emphasize Q3's roll as rectifier.
Q3 and Q4 are a current mirror.
As the current in Q3's base emitter junction increases an identical mirrored current is setup in the collector of Q4.

The action of the current mirror serves to invert polarity for positive inputs.
When the collector currents of Q2 and Q4 are summed the final result provides the absolute value.
The output of the current rectifier is a current that can be directly connected to the RMS stage.

Q3 and Q4 require matched devices in thermal equilibrium.
For a stereo detector a single THAT300 could be used and share between channels.

Advantages and Disadvantages.

The diode-base absolute value circuit has about 10 dB or so less dynamic range but does not require matched transistors in the rectifier.
The current rectifier has more dynamic range but requires matched devices.

I may have missed a few subtleties but really wanted to get this posted before this session times out. LOL.

Added degeneration resistors in series with the current mirror device emitters would reduce the sensitivity to Vbe matching.

Being able to capacitor couple the input resistor eliminates input offset DC performance errors affecting low level accuracy/dynamic range. (perhaps not a concern with better modern op amps.).

JR

[mediatechnology]

While that looks like the 2252 it lacks some typical RMS aspects. The classic RMS math is square root of the integral of signal squared.

True.



Since the VCAs being used with this detector presumably have a 6 mV/dB scale factor (THAT218X) the resulting response is RMS.
The square root is effectively taken downstream from the detector in the VCA by virtue of it's 6mV/dB response.
But the RMS representation, at 6mV/dB is still in the log domain.
If the audio input of the VCA were a fixed DC reference, the VCA output would represent the DC RMS value in the linear domain.
That would be useful in a measurement instrument.

I won't be a purist or pedant about true or real RMS vs ave because my development work while developing my last peak/VU console meters revealed very little difference

People seem to prefer the way RMS sounds. I'm not going to argue with success.

Related reading:
Compressor Attack Release Signature Comparisons viewtopic.php?f=6&t=281

[JR.]

We've been around that tree before, I think we discussed this exactly in another thread about RMS. DBX also made a big deal about RMS being special for tape NR a much bigger business for them than dynamics back in the day (they claimed RMS introduced less companding errors from tape's less than ideal transfer function).

2xVbe modulated by audio signal current is indeed 2x LOGx or x^2. Then subtracting two constant current biased Vbe is not remotely dividing it by 2, or taking a square root. It gives the correct divide by 2 answer at 0dB, while it is probably a more useful gain law for their VCAs (-6mV/dB).

The charging current at Ct is arguably X^2-(RT/15V) but discharge current is simply -RT/15V for a roughly linear -dB/sec decay (for useful range of the side chain voltage). I suspect that makes a useful sounding fast attack/slow release side chain characteristic.

If it sounds good it is good, enjoy.

Sorry, disregard the grumpy old man...

JR